1. Field
Embodiments of the invention relate to electronic devices, and more particularly, in one or more embodiments, to radio frequency receivers.
2. Description of the Related Technology
Many electronic systems operate with radio-frequency (RF) signals. Such electronic systems can include an RF receiver that processes a wireless or wired signal (for example, a radio frequency signal) received via a wireless medium, such as air, or over a wire, such as a copper cable. An RF receiver can include various components to amplify and/or filter an RF signal to recover original data carried by the RF signal.
Referring to FIG. 1, a conventional direct-conversion radio frequency (RF) receiver system will be described below. The illustrated system 100 includes an antenna 101, an input stage structure 110, an input matching network 120, a low noise amplifier (LNA) 130, a first mixer 140a, a second mixer 140b, a local oscillator 142, a phase shifter 144, a first low pass filter (LPF) 150a, a second low pass filter 150b, a first analog-to-digital converter (ADC) 160a, a second analog-to-digital converter (ADC) 160b, an adaptive I/Q compensation filter 170, an adder 180, and a baseband module 190. The adaptive I/Q compensation filter 170 and the adder 180 can be collectively referred to as an “I/Q compensation module” or “I/Q compensation block” in the context of this document. In an instance in which the receiver system is used for processing a wired signal, the antenna 101 can be omitted.
Since there are two mixers operating with 90° phase offset, the RF receiver system 100 can be referred to as using “quadrature” reception. The PQ compensation module can be used with super-heterodyne receivers, or any other RF receiver that employs quadrature reception, even if this quadrature operation occurs in only one stage of the RF receiver.
The antenna 101 is configured to receive an RF signal. The antenna 101 can be any suitable antenna for wireless signal reception. The antenna 101 provides the received wireless signal to the input stage structure 110.
The input stage structure 110 serves to receive and process the RF signal. The input stage structure 110 can include, for example, an antenna interface circuit to interface with the antenna 101, and a filter (for example, a band pass filter) to filter out signals outside of a signal band of interest. The input stage structure 110 generates a first processed signal, which is provided as an input to the input matching network 120.
The input matching network 120 serves to improve power transfer from the input stage structure 110 to the low noise amplifier 130, and to reduce signal reflection from the low noise amplifier 130. Further, the input matching network 120 can serve to improve the noise performance of the low noise amplifier 130. The input matching network 120 is configured to match the impedance of the low noise amplifier 130 with the impedance of the structure (for example, the input stage structure 110 and the antenna 101) on the opposite side of the input matching network 120 from the low noise amplifier 130. The input matching network 120 receives the first processed signal from the input stage structure 110, and generates a second processed signal z(t), which is provided as an input to the low noise amplifier 130.
The low noise amplifier 130 serves to amplify the second processed signal z(t) from the input matching network 120 to generate an amplified signal, and provides the amplified signal to the first and second mixers 140a, 140b. The low noise amplifier 130 is configured to amplify a relatively weak signal with a gain such that the effect of noise on subsequent stages of the receiver system 100 is reduced.
The first mixer 140a serves to mix the amplified signal from the low noise amplifier 130 and a first local frequency signal LOI from the phase shifter 144 to generate a first mixed signal. The first mixed signal can include the fundamental frequencies of the current signal, the first local frequency signal, harmonics thereof, and intermodulation products. The second mixer 140b serves to mix the amplified signal from the low noise amplifier 130 and a second local frequency signal LOQ from the phase shifter 144 to generate a second mixed signal. The second mixed signal can include the fundamental frequencies of the current signal, the second local frequency signal, harmonics thereof, and intermodulation products.
In the illustrated example, the first local frequency signal LOI can be used to process in-phase (I) components of the received signal while the second local frequency signal LOQ can be used to process quadrature (Q) components of the received signal. Ideally, the first and second local frequency signals LOI, LOQ should have a phase difference of 90 degrees from each other. The phase shifter 144 is configured to generate such a phase difference, using a local oscillation signal from the local oscillator 142. These components can also exist in other types of RF receivers, such as super-heterodyne or low-IF receivers.
The first and second low pass filters 150a, 150b serve to filter the first and second mixed signals, respectively, and provide the filtered mixed signals to the first and second analog-to-digital converters 160a, 160b, respectively. The first and second low pass filters 150a, 150b are for anti-aliasing, and pass frequencies up to some cut-off frequency. These filters block higher frequencies beyond this cut-off frequency.
The first and second analog-to-digital converters 160a, 160b serve to convert the filtered mixed signals from analog form into a digital signal x[n]. The first mixer 140a, the first LPF 150a, and the first ADC 160a form an in-phase or I path. The second mixer 140b, the second LPF 150b, and the second ADC 160b form a quadrature-phase or Q path. The first and second analog-to-digital converters 160a, 160b can provide the digital signal x[n] as an input to the adaptive filter 170 and the adder 180. The output of the first ADC 160a forms the real number portion of the digital signal x[n], and the output of the second ADC 160b forms the imaginary number portion of the digital signal x[n].
The adaptive filter 170 is configured to generate a compensation signal to compensate for an imbalance between the I path and the Q path. Such an imbalance can be referred to as “I/Q imbalance” or “I/Q mismatch” in the context of this document, and will be described later in detail. The adaptive filter 170 can use a feedback signal from the adder 180, and the digital signal x[n] to generate the compensation signal that is to be provided to the adder 180.
The adder 180 is configured to add the digital signal x[n] and the compensation signal, and provides the compensated signal to the baseband module 190. The baseband module 190 receives the compensated signal from the adder 180, and performs digital signal processing on the signal. The digital signal processing can include, for example, demultiplexing and decoding.
In an RF receiver such as a direct-conversion receiver, I/Q imbalance occurs, for example, when the transfer function of the I path of the receiver is different from that of the Q path of the receiver, and/or when the phase relationship between the two paths is not quite 90 degrees. Such imbalance occurs due to imperfections and variability of the analog components of an RF receiver, such as the filters, mixers, amplifiers, and ADCs. Sources of such imbalances include, but are not limited to, gain and phase mismatch of the mixers, frequency responses of low pass filters, gain and offset of ADCs, ADC-clock timing mismatch, and a non-linear I/Q imbalance. I/Q imbalance is typically unavoidable using state-of-art analog circuit implementations.
I/Q imbalance can adversely affect the performance of an RF receiver. For example, I/Q imbalance can decrease the image rejection ratio (IRR) of an RF receiver down to, for example, 20-40 dB, resulting in crosstalk or interference between mirror frequencies. Thus, I/Q imbalance reduces the signal-to-noise ratio of the receiver 100, and increase the number of bit errors for a given data rate. Thus, I/Q imbalance needs to be reduced or cancelled.
I/Q imbalance can produce an undesired image signal, which falls within the band of interest. The term “image signal” refers to an undesired signal at frequencies occupied by the desired input signal. I/Q imbalance is a potential source of interference to proper reception. The term “image rejection ratio” is a measure of image strength relative to desired signal, and can refer to a ratio of (a) power of the desired signal to (b) power of the image signal. The image rejection ratio is usually expressed in decibels (dB). A desired IRR performance can be at least 45 dB, for example, in cable-modem applications where the desired baseband signal occupies 50-70 MHz of bandwidth.
There have been various attempts to reduce or eliminate I/Q balance from RF receivers. Among others, digital signal processing techniques have been used to reduce I/Q imbalance. Some of these techniques focus on frequency-independent I/Q imbalance compensation in specific architectures and assume certain modulation schemes possibly combined with some known pilot or training data. Other techniques attempt to compensate for frequency-dependant imbalances, assuming known pilot data or using interference cancellation (IC) or blind signal separation (BSS) principles.
Among the techniques for frequency-dependent I/Q imbalance compensation, Anttila et al., “Circularity-Based I/Q Imbalance Compensation in Wideband Direct-Conversion Receivers,” IEEE Transactions on Vehicular Technology, Vol. 57, No. 4, pp. 2099-2113 (July 2008), presents a blind (non-data-aided) circularity-based compensation of frequency-dependent I/Q imbalances in RF receivers.
Referring to FIG. 2A, an I/Q compensation module disclosed by Anttila et al. will be described below. The illustrated I/Q compensation module 200 includes a first node 201, a second node 202, a complex conjugation block 210, an adaptive filter 220, a delay element 225, a filter adaptation block 227, and an adder 230. A digital signal x[n] is provided to the first node 201 from the I and Q paths of a receiver, such as the I and Q paths of FIG. 1. The digital signal x[n] is provided to the adder 230 and the conjugation block 210.
In the context of this document, “n” denotes a discrete-time index, where the time-interval between indices can be found from the sampling rate. In the context of this document, a discrete-time sequence of samples “x” is referred to as “x[n].” For simplicity of notation, “x[n]” can also be used to indicate the value of the sequence “x” at time-index “n.” Vectors in boldface, such as xn, will be used to refer to the vector at time-index “n.”
The conjugation block 210 is configured to change the polarity of the imaginary number part of the digital signal x[n], thereby generating a complex conjugate signal x*[n] of the digital signal x[n]. For example, the digital signal x[n] can be a sequence of complex numbers. For example, let one sample in the sequence be expressed as a+jb, in which a is the real number part, jb is the imaginary number part, and j corresponds to the square root of −1. The complex conjugate x*[n] has the same real part and has an imaginary part having the same magnitude and the opposite sign. In the example above, the complex conjugate can be expressed as a−jb.
The adaptive filter 220 can be a finite impulse response (FIR) filter, whose coefficients at time-index “n” can be expressed as the vector wn. The FIR filter is a type of a discrete-time filter. The FIR filter can generate an output digital sequence v[n] as expressed in Equation 1 below.v[n]=wn[0]x[n−N+1]+wn[1]x[n−N]+ . . . +wn[i]x[n−N+i]+ . . . +wn[N−1]x[n]  Equation 1
In Equation 1, variable x[n] is the input signal, and variable v[n] is the output signal. Weights wn[i], (i=0, 1, 2, . . . , N−1) are filter coefficients at time “n,” also known as tap weights. N is the filter order or length, and an (N+1)th order filter has N terms, each of which can be referred to as a tap. For example, weight wn[0] can be referred to as a first tap at time “n,” which can correspond to a tap that is not delayed. Weight wn[1] can be referred to as a second tap. wn[N−1] can be referred to as an N-th tap.
The adaptation filter 220 receives a feedback signal λ yn y[n] from the adder 230 via the delay element 225 and the filter adaptation block 227. In the illustrated example, an output signal y[n] from the adder 230 is delayed by the delay element 225. The amount of a delay provided by the delay element 225 can be at least one sample. The delayed output signal is provided to the filter adaptation block 227, which generates the feedback signal λ yn y[n]. The feedback signal λ yn y[n] is used by the adaptive filter 220 to generate a compensation signal to be added to the input signal x[n] at the adder 230 to cancel or reduce I/Q imbalance.
The adaptive filter 220 is configured to perform a complex-convolution operation (*) on the conjugate signal x*[n] with the adaptive filter signal wn to generate a compensation signal expressed as x*[n]*wn. The convolution operation can be computed as the sum of the product of the two sequences after one is reversed and shifted on the time axis. The adaptive filter wn can be iteratively updated as in Equation 2 below.wn+1=wn−λyny[n]  Equation 2
In Equation 2, y[n] is the compensated signal value at time “n.” This value can be expressed as y[n]=x[n]+v[n]=wnTxn, wherein wn Δ [wn[0], wn[1], wn[2], . . . , wn[N−1]]T denotes the vector of coefficients (alternatively “filter coefficients”) of the compensator at time index n, and the vector xnΔ[x[n−N+1], x[n−N], . . . , x[n]]T. λ denotes the adaptation step size or adaptation rate, and ynΔ [y[n], y[n−1], . . . , y[n−N+1]]T.
The adder 230 adds the compensation signal to the digital signal x[n], and provides the compensated signal y[n] to the baseband module 190 (FIG. 1) of the receiver. The compensated signal sequence y[n] is provided as a feedback signal to the adaptive filter 220 through the delay element 225 and the filter adaptation block 227.
The technique described in the above example is blind to the received baseband signal, which is useful since a training signal need not be applied at the receiver input. The technique is adaptive in that it accounts for time-varying mismatches. The technique is also frequency-selective, and thus can be suitable for correction of wideband channels.
Equation 2, above, indicates that the technique determining filter coefficients wn blindly (i.e., without knowledge of the baseband signal) and adaptively by employing the signal “properness” property, that is, determining if y[n] is a proper signal sequence, which indicates that y[n] and y*[n] are uncorrelated. In the technique of Anttila et al., λ is a fixed training coefficient selected by the user. After convergence, the signal y[n] should exhibit the condition expressed in Equation 3 below, in which E denotes expected value operator (or expectation operator) that provides the long-run average.E[y[n−i],y[n]]=0,where i=0,1, . . . N−1.  Equation 3